Display device with transistor sampling for improved performance

ABSTRACT

A display device including a pixel array having a plurality of pixels arranged in a matrix form, each of the pixels including a sampling transistor configured to sample a data potential from a video signal line which is insulated from and intersects a control line in response to the change in potential of the control line, and a light-emitting element configured to emit light at the brightness commensurate with the magnitude of the post-sampling data potential.

RELATED APPLICATION DATA

This application is a continuation of U.S. patent application Ser. No.14/557,917 filed Dec. 2, 2014, which is a continuation of U.S. patentapplication Ser. No. 13/742,525 filed Jan. 16, 2013, now U.S. Pat. No.8,941,564 issued on Jan. 27, 2015 which is a continuation of U.S. patentapplication Ser. No. 12/464,534 filed May 12, 2009, now U.S. Pat. No.8,390,540 issued Mar. 5, 2013 the entireties of which are incorporatedherein by reference to the extent permitted by law. The presentapplication also claims priority to Japanese Priority Patent ApplicationNo. JP2008-126118 filed May 13, 2008, the entirety of which isincorporated by reference herein to the extent permitted by law.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a display device which includes a pixelarray having a plurality of pixels arranged in a matrix form andwirings, common to the plurality of pixels and disposed in twodirections, intersecting with each other in the pixels. Morespecifically, the present invention relates to a wiring formingtechnique for minimizing the number of wirings disposed in the twodirections and bringing the resistances thereof close to each other tothe extent possible.

2. Description of the Related Art

Some display devices use electro-optical elements whose brightnesschanges with change in voltage applied thereto or current flowingtherethrough. For example, a liquid crystal display element is a typicalexample of an electro-optical element whose brightness changes withchange in voltage applied thereto. A display device using such elementsis called a “liquid crystal display device.”

On the other hand, an organic electro-luminescence element is a typicalexample of an electro-optical element whose brightness changes withchange in current flowing therethrough. An organic electro-luminescenceelement is commonly called an OLED (Organic Light Emitting Diode). Aliquid crystal display element is an optical modulation element (i.e.,non-self-luminous) adapted to modulate light from the light source. AnOLED differs from a liquid crystal display element in that the former isa self-luminous element which can emit light by itself. A display deviceusing OLEDs will be herein referred to as an “organic EL displaydevice.”

Such liquid crystal display devices have become increasingly large inscreen size, high in definition and low in cost, finding widespread useon a full-scale basis ahead of a complete transfer to digitalbroadcasting. As a result, these display devices are demanded to befurther reduced in cost.

On the other hand, organic EL display devices offer many advantages overtheir liquid crystal counterparts, including no need for backlight,lower power consumption and possible reduction in thickness, andexcellent moving image display performance. As a result, these devicesare seen as a next-generation of display devices. Development effortsare moving at an accelerated pace to replace some of the liquid crystaldisplay devices with organic EL ones with an eye toward widespread useof liquid crystal display devices on a full-scale basis.

In the development efforts for display devices typified by liquidcrystal and organic EL display devices, it is essential that themanufacturing process be as simple as possible to prevent point and linedefects for enhanced yield in order to provide a larger screen size. Inparticular, the empirical rule says that the fewer the lithographyprocess steps adapted to perform patterning using resist, an organicmaterial, the significantly less likely that defects will occur. Basedon this rule, the wiring process is undergoing simplification (refer,for example, to a wiring structure having one aluminum layer describedin Japanese Patent Laid-Open No. 2004-4192 and Japanese Patent Laid-OpenNo. 2004-317916 (hereinafter referred to as Patent Documents 1 and 2regarding organic EL display device).

Both of the techniques disclosed in Patent Documents 1 and 2 offer asimpler wiring structure in the pixel using chromium layer and itsoverlying layer. The chromium layer is at the same hierarchical leveland made of the same material as the gate electrode of the thin filmtransistor (TFT). The overlying layer is made of aluminum.

SUMMARY OF THE INVENTION

In the wiring structure described in Patent Document 1, no contacts areprovided midway along the control lines for the sampling transistors andholding capacitors or along the video single lines and source voltagesupply lines. The control lines are disposed in the row direction in thepixel (horizontally on the display screen). The video single lines andsource voltage supply lines are disposed in the column direction(vertically on the display screen).

This prevents reduction in potential resulting from the contactresistance.

In the wiring structure described in Patent Document 1, however, all thecontrol lines in the row direction are formed with a chromium layer,i.e., the gate metal. As a result, a large screen leads to a largesignal delay in the control lines running in the row direction, readilyresulting in a deviation in sampling timing particularly between pixelsin the same row close to and far from the vertical drive circuit adaptedto control the control lines. A large deviation in sampling timing givesrise to degraded display quality due to sampling of the video signal ata wrong level unless corrected, for example, by securing a large marginfor variation of the video signal. On the other hand, the increasedmargin inhibits high-speed display. In both cases, therefore, the wiringstructure described in Patent Document 1 is disadvantageous in that itis difficult to achieve larger screen size and higher definition.

The wiring structure described in Patent Document 2 has the samedisadvantage as that described in Patent Document 1 because all thecontrol lines in the row direction are formed with a chromium layer,i.e., the gate metal.

Further, in the wiring structure described in Patent Document 2, thesignal line and power source supply line in the column direction areadjacent to each other between the adjacent pixels. Moreover, the twolines are formed with an aluminum layer at the same hierarchical level.Therefore, part of one of the two wirings adjacent to each other in thecolumn direction has a lower bridge structure made of the same chromiumlayer, i.e., the gate metal, so as to reduce the probability of shortcircuit between the adjacent wirings due to inclusion of foreignobjects. It should be noted, however, that because the wirings in therow direction are formed with the same chromium layer, the intersectionsbetween the wirings in the row and column directions cannot have thislower bridge structure. Therefore, the wiring structure described inPatent Document 2 is only limitedly effective in preventing shortcircuit between adjacent wirings.

Still further, the wiring structure described in Patent Document 2 hasother disadvantage in addition to the signal delay in the wirings in therow direction. That is, the wirings in the column direction arealternately made of aluminum and chromium layers, and the two wiringsare connected by a single contact hole. This leads to not only increasedresistance but also signal delay due to contact connection resistance inthe wirings in the column direction. If the video signals in the columndirection have a large delay when there is already a rather large delayin the control signals in the row direction, it will be increasinglydifficult to design the sampling timings properly, thus making it evenmore difficult to achieve larger screen size and higher definition.

A possible comprehensive solution to all the disadvantages of thebackground technology described above would be to provide an additionalaluminum layer. However, this leads to increased manufacturing cost,making this solution unpractical.

It should be noted that other types of display devices such as liquidcrystal display device have the same problems (difficulties in achievinglarger screen size and higher definition) although no specific patentpublication numbers are given. However, the proper video signal samplingtimings are more essential for an organic EL display device than anyother type of display device. Therefore, these difficulties are a moreserious problem for organic EL display devices.

A display device according to an embodiment (first embodiment) of thepresent invention relates to a display device such as so-called organicEL display device having light-emitting elements. That is, the displaydevice according to this embodiment includes a pixel array having aplurality of pixels arranged in a matrix form. Each of the plurality ofpixels includes a sampling transistor and light-emitting element. Thesampling transistor samples a data potential from a video signal linewhich is insulated from and intersects a control line in response to thechange in potential of the control line. The light-emitting elementemits light at the brightness commensurate with the magnitude of thepost-sampling data potential.

Further, in the above display device, one of the video signal line andcontrol line includes a high-fusion-point metal wiring layer andoverlying wiring layer. The high-fusion-point metal wiring layer is madeof the same material as the gate of the sampling transistor. Theoverlying wiring layer is connected to the high-fusion-point metalwiring layer via a contact hole. The other of the video signal line andcontrol line is a single layer wiring provided at the same hierarchicallevel and made of the same material as the overlying wiring layer whichintersects the one of the video signal line and control line at aportion of the layer which includes the high-fusion-point metal wiringlayer. In the one of the video signal line and control line, theplurality of contact holes are provided for each wiring connectionsection adapted to connect an end portion of the layer including thehigh-fusion-point metal wiring and an end portion of the overlyingwiring layer.

A display device according to another embodiment (second embodiment) ofthe present invention has the following feature in addition to thefeatures according to the first embodiment. That is, the layer includingthe high-fusion-point metal wiring contains a metallic material having ahigher wiring resistance and higher fusion point than the overlyingwiring layer.

A display device according to still another embodiment (thirdembodiment) of the present invention has the following feature inaddition to the features according to the first embodiment. That is,when the number of the contact holes is increased for each of the wiringconnection sections, only a given number of the contact holes areprovided if a signal waveform of the video signal line no longer showsany change of response with the given number of the contact holes ormore provided.

A display device according to still another embodiment (fourthembodiment) of the present invention has the following feature inaddition to the features according to the first embodiment. That is, asource voltage supply line adapted to pass a drive current through thelight-emitting element is disposed in the pixel in parallel with thecontrol line. The intersection between the source voltage supply lineand video signal line is formed with other high-fusion-point metalwiring layer provided at the same hierarchical level and made of thesame material as the layer including the high-fusion-point metal wiring.As many contact holes as in the wiring connection section are formed inother wiring connection section between the other layer including thehigh-fusion-point metal wiring and the overlying wiring layer.

Further preferably, the layer including the high-fusion-point metalwiring and the other layer including the high-fusion-point metal wiringare connected so as to form a single pattern (fifth embodiment).

Further preferably, the overlying wiring layer is disposed so as to forma length while at the same time providing a predetermined spacing withthe control line and source voltage supply line respectively to preventshort circuit. The contact holes shared by the wiring connection sectionand other wiring connection section are arranged along the length of theoverlying wiring layer (sixth embodiment). Further preferably, as manyof the contact holes as possible are arranged over the entire lengththereof.

A display device according to still another embodiment (seventhembodiment) of the present invention has the following feature inaddition to the features according to the first embodiment. That is, thenumber of the contact holes of the wiring connection section is largerthan the maximum number of contact holes provided for each of the otherconnection sections in the pixel, including the connection sectionadapted to connect the control line to the control node of the samplingtransistor.

A display device according to still another embodiment (eighthembodiment) of the present invention includes a pixel array having aplurality of pixels arranged in a matrix form. First and second wiringsare provided at least one each for each of the plurality of pixels. Thefirst wiring is disposed in one direction. The second wiring is disposedin a direction orthogonal to the first wiring. One of the first andsecond wirings is formed with a high-fusion-point metal wiring layer,contact hole and overlying wiring layer. The contact hole is provided inan insulating layer on the high-fusion-point metal wiring layer. Theoverlying wiring layer is connected to the high-fusion-point metalwiring layer via the contact hole and provided on the insulating layer.The other of the first and second wirings is a single layer wiringprovided at the same hierarchical level and made of the same material asthe overlying wiring layer. The plurality of contact holes are providedfor each wiring connection section adapted to connect an end portion ofthe layer including the high-fusion-point metal wiring and an endportion of the overlying wiring layer.

Further preferably, the layer including the high-fusion-point metalwiring contains a metallic material having a higher wiring resistanceand higher fusion point than the overlying wiring layer (ninthembodiment).

The display devices according to the above-described first to seventhembodiments will be described below. In this description, we assume, asan example, that the video signal line has a two-layered wiringstructure in which the layer including the high-fusion-point metalwiring and the overlying wiring layer are connected by a plurality ofcontacts, that the control line (and source voltage supply line) areeach a single layer wiring provided at the same hierarchical level andmade of the same material as the overlying wiring layer, and that thecontrol line (and source voltage supply line) are disposed in adirection different from that of the video signal line. The presentinvention also includes the case in which the terms “video signal line”and “control line” are interchanged in the following description.

First, the control line is formed, for example, as a single wiring layerwith the same material as the overlying wiring layer which is lower inresistance than the layer including the high-fusion-point metal wiringmade of the same material as the gate of the sampling transistor.Because of lack of contacts midway along its length, the control linehas a smaller signal delay than if the same line was formed with ahigh-fusion-point metal. Here, the term “(signal) delay amount” is acomprehensive concept which includes degraded, if not delayed, responseof the signal waveform (the same holds true for the following functionsand embodiments described layer).

On the other hand, the video signal line has the above-describedtwo-layered wiring structure. Thanks to the plurality of contact holesprovided for each of the wiring connection sections between the layerincluding the high-fusion-point metal and the overlying wiring layer,the delay amount of the video signal is smaller than if there was onlyone contact hole.

Normally, when the activation timing (sampling start timing) of thecontrol signal for the sampling transistor is set during a period oftime in which the video signal pulse is maintained at the data voltagelevel after making a transition from the inactive level, the samplingstart timing is designed to include a predetermined time margin from theabove transition in the video signal pulse level.

However, if the signal delay amounts of both of the control line andvideo signal lines are large, it is difficult to achieve proper timingsin the two wiring directions.

In the first to seventh embodiments of the present invention, one of thewirings (wiring direction of the control line) which intersect eachother is a single layer wiring formed with the overlying wiring layerwhich has a small delay amount thanks to its small resistance. Further,the other wiring (video signal line) has the plurality of contacts toprovide a reduced delay amount. This prevents sampling of incorrectlevels even if the margin is set smaller in the timing design.Conversely, an increased margin prevents video quality degradationresulting from sampling of incorrect levels.

This makes all the easier a display speedup (higher-frequency driving)associated with larger screen size and higher definition.

Although relating to display devices including not only self-luminousones such as organic EL display devices but also optical modulation onessuch as liquid crystal display devices, the display devices according tothe eighth and ninth embodiments provide almost the same functions asdescribed above because these devices demand video signal sampling.

The present invention permits easy design of the sampling timing forvarious types of display devices including self-luminous ones such asorganic EL display devices and optical modulation ones such as liquidcrystal display devices, thus alleviating the difficulties involved inachieving larger screen size and higher definition and at the same timeproviding low-cost display devices.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating the major components of an organic ELdisplay device according to an embodiment of the present invention;

FIG. 2 is a basic configuration diagram of a pixel circuit 1 accordingto the embodiment of the present invention;

FIG. 3 is a basic configuration diagram of a pixel circuit 2 accordingto the embodiment of the present invention;

FIG. 4 is a graph and equation illustrating the characteristic of anorganic light-emitting diode;

FIG. 5 is a basic configuration diagram of a pixel circuit 3 accordingto the embodiment of the present invention;

FIGS. 6A to 6I are timing diagrams illustrating the waveforms of varioussignals and voltages during display control according to the embodimentof the present invention;

FIGS. 7A to 7C are diagrams describing the operation until sampling;

FIGS. 8A to 8C are diagrams describing the operation until a secondthreshold correction;

FIGS. 9A and 9B are diagrams describing the operation until a thirdthreshold correction;

FIG. 10 is a graph illustrating the change in source potential over timeaccording to the embodiment of the present invention;

FIGS. 11A to 11C are diagrams describing the operation until a lightemission period;

FIGS. 12A and 12B are plan views of a pixel circuit according to theembodiment of the present invention;

FIGS. 13A and 13B are sectional views of the pixel circuit according tothe embodiment of the present invention; and

FIG. 14 is a sectional view of the pixel circuit according to amodification example of the embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention may be applied to liquid crystal and other displaydevices. However, embodiments of the present invention will be describedbelow with reference to the accompanying drawings taking, as an example,a case in which the present invention is applied to an organic ELdisplay device.

<Overall Configuration>

FIG. 1 is a diagram illustrating the major components of an organic ELdisplay device according to an embodiment of the present invention.

An organic EL display device 1 illustrated in FIG. 1 includes a pixelarray 2 and drive circuit. The pixel array 2 has a plurality of pixelcircuits 3 (i,j) arranged in a matrix form. The drive circuit drives thepixel array 2 and includes a vertical drive circuit (V scanner) 4 andhorizontal drive circuit (H scanner or H. Scan).

The plurality of V scanners 4 are provided depending on theconfiguration of the pixel circuits 3. Here, the V scanner 4 includes ahorizontal pixel line drive circuit (Drive Scan) 41 and write signalscan circuit (Write Scan) 42.

Reference numeral 3 (i,j) of the pixel circuits shown in FIG. 1 denotesthat each pixel circuit has a vertical address i (I=1 or 2) andhorizontal address j (j=1, 2 or 3). Each of these addresses i and j takeon an integer equal to or greater than 1. The maximum values of i and jare “n” and “m,” respectively. Here, for simplification of the diagram,a case will be shown in which n=2 and m=3.

This address notation is similarly applied to the elements, signals,signal lines, voltages and others of the pixel circuits in thedescription and drawings which follow.

Pixel circuits 3(1,1) and 3(2,1) are connected to a common verticalfirst signal line SIG(1). Similarly, pixel circuits 3(1,2) and 3(2,2)are connected to a common vertical second signal line SIG(2). Stillsimilarly, pixel circuits 3(1,3) and 3(2,3) are connected to a commonvertical third signal line SIG(3).

A first scan signal VSCAN1(1) can be applied to the pixel circuits3(1,1), 3(1,2) and 3(1,3) in the first row from the horizontal pixelline drive circuit 41 via a common scan signal line. Similarly, a firstscan signal VSCAN1(2) can be applied to the pixel circuits 3(2,1),3(2,2) and 3(2,3) in the second row from the horizontal pixel line drivecircuit 41 via a common scan signal line.

Further, a second scan signal VSCAN2(1) can be applied to the pixelcircuits 3(1,1), 3(1,2) and 3(1,3) in the first row from the writesignal scan circuit 42 via other common scan signal line. Similarly, asecond scan signal VSCAN2(2) can be applied to the pixel circuits3(2,1), 3(2,2) and 3(2,3) in the second row from the write signal scancircuit 42 via other common scan signal line.

<Pixel Circuit 1>

FIG. 2 illustrates the most basic configuration of the pixel circuit3(i,j) when the drive transistor includes a p-channel metal oxidesemiconductor (PMOS) transistor.

The pixel circuit 3(i,j) illustrated in FIG. 2 controls an organiclight-emitting diode OLED serving as a light-emitting element. The pixelcircuit includes a drive transistor Md, sampling transistor Ms andholding capacitor Cs, in addition to the organic light-emitting diodeOLED. The drive transistor Md includes a PMOS TFT. The samplingtransistor Ms includes an NMOS TFT.

Although not specifically illustrated, the organic light-emitting diodeOLED has a layered body with a second electrode (cathode electrode)formed thereon. The layered body makes up an organic film. This body isformed by sequentially depositing, on a substrate made, for example, oftransparent glass, a first electrode (anode electrode), holetransporting layer, light-emitting layer, electron transporting layer,electron injection layer and other layers. The anode electrode isconnected to a positive first power source, and the cathode electrode toa negative second power source. It should be noted that the anodeelectrode may be positive, and the cathode electrode negative. In thiscase, the anode electrode is connected to the second power source, andthe cathode electrode to the first power source.

It should be noted that FIG. 2 illustrates a case in which the anode ofthe organic light-emitting diode OLED is supplied with a source voltageVDD from the positive first power source, and that the cathode thereofis connected to a reference voltage such as ground voltage GND.

When a predetermined bias voltage is applied between the cathode andanode of the organic light-emitting diode OLED, the injected electronsand holes recombine in the light-emitting layer, thus emitting light.Because the organic light-emitting diode OLED can emit any of red (R),green (G) and blue (B) lights by proper selection of the organicmaterial making up the organic film, the same diode OLED is capable ofdisplaying a color image, for example, if the pixels are arranged ineach row in such a manner that R, G and B lights are emitted.Alternatively, a white light-emitting organic material may be used, withthe distinction between R, G and B made using filter colors. Stillalternatively, four colors, namely, R, G, B and W (white), may be usedinstead.

The drive transistor Md functions as a current control section adaptedto control the amount of current flowing through the organiclight-emitting diode OLED so as to determine the display gray level.

The drive transistor Md has its source connected to the supply line ofthe source voltage VDD and its drain connected to the anode of theorganic light-emitting diode OLED.

The sampling transistor Ms is connected between a supply line (signalline SIG(j)) of a data potential Vsig and the gate (control node NDc) ofthe drive transistor Md. The data potential Vsig determines the pixelgray level. The same transistor Ms has one of its source and drainconnected to the gate of the drive transistor Md and the other thereofconnected to the video signal line SIG(j). The data potential Vsig isapplied to the signal line SIG(j) from the H scanner 5. The samplingtransistor Ms samples the data having the level to be displayed by thepixel circuit at a proper timing during this data potential applicationperiod. This is done to eliminate the adverse impact of unstable levelduring the transition period on the display image.

The holding capacitor Cs is connected between the supply line of thesource voltage VDD and the gate of the drive transistor Md. The roles ofthe holding capacitor Cs will be described later in relation to theoperation.

It should be noted that the components controlled by the horizontalpixel line drive circuit 41 are omitted in FIG. 2. An example of suchcomponents may be, for example, another transistor connected between thesupply line of the source voltage VDD and the gate of the drivetransistor Md. Alternatively, another example of such components may bethat adapted to repeatedly apply the source voltage VDD for apredetermined time at constant intervals. Such components are providedfor purposes of drive scan. However, they are omitted in FIG. 2 becauseof many schemes available for drive scan.

<Pixel Circuit 2>

FIG. 3 illustrates the most basic configuration of the pixel circuit3(i,j) when the drive transistor includes an N-channel metal oxidesemiconductor (NMOS) transistor.

The pixel circuit 3(i,j) illustrated in FIG. 3 is similar to that inFIG. 2 except that the drive transistor Md is of different channelconductivity type from the one shown in FIG. 2. Using an NMOS transistoras the drive transistor Md offers two advantages. Firstly, a large drivecurrent can be used per unit size. Secondly, N-channel transistors canbe used for all the transistors in the pixel circuit, thus providing asimpler manufacturing process.

It should be noted that all transistors in the pixel circuits 1 and 2are formed by TFTs. The thin film semiconductor layer used to form theTFTs is made of a semiconductor material such as polysilicon oramorphous silicon. Polysilicon TFTs can have a high mobility but varysignificantly in their characteristics, which makes these TFTs unfit forachieving a larger screen size of the display device. Therefore,amorphous TFTs are generally used in a display device having a largescreen. It should be noted, however, that P-channel TFTs are difficultto form with amorphous silicon TFTs. As a result, the pixel circuit 2 ora pixel circuit based thereon should preferably be used.

Here, the above-described pixel circuits 1 and 2 are examples of a pixelcircuit applicable to the present embodiment, i.e., basic examples of atwo-transistor (2T) one-capacitor (1C) configuration. Therefore, thepixel circuit which can be used in the present embodiment may have anadditional transistor and/or capacitor in addition to the pixel circuit1 or 2 as a basic configuration. More specifically, the pixel circuitapplicable to the present embodiment may have, for example, any of 4T1C,4T2C and 5T1C pixel circuits, though detailed description is omitted.

<Outline of Light Emission Control>

The light emission control operation of the above two pixel circuits canbe briefly described as follows.

The holding capacitor Cs is coupled to a control node NDc of the drivetransistor Md. The signal voltage Vsig from the signal line SIG(j) issampled by the sampling transistor Ms. The obtained data potential Vsigis applied to the control node NDc.

FIG. 4 illustrates the I-V characteristic graph of the organiclight-emitting diode OLED and a general equation of a drain current Idsof the drive transistor Md (corresponds to a drive current Id of theOLED).

When the predetermined data potential Vsig is applied to the gate of thedrive transistor Md, the P-channel drive transistor Md in the case ofthe pixel 1 shown in FIG. 2 is designed to operate in the saturationregion at all times with its source connected to the power source. As aresult, the P-channel drive transistor Md functions as a constantcurrent source having a current level denoted by the equation shown inFIG. 4. The drain current Ids supplied by this constant current sourceis determined by a gate-to-source voltage Vgs whose level iscommensurate with the data potential applied to the gate of theP-channel drive transistor Md. Therefore, the organic light-emittingdiode OLED emits light at the brightness commensurate with thepost-sampling data potential Vsig.

As is well known, the I-V characteristic of the organic light-emittingdiode OLED changes as illustrated in FIG. 4 due to secular change. Atthis time, the constant current source attempts to supply the same levelof the drive current Id. This increases a voltage V applied to theorganic light-emitting diode OLED, pushing up the drain potential of theP-channel drive transistor Md. However, the gate-to-source voltage Vgsof the same transistor Md remains constant. Therefore, the constantdrive current Id flows through the same diode OLED. As a result, thelight emission brightness remains unchanged.

In the case of the pixel 2 shown in FIG. 3 having an N-channeltransistor as the drive transistor Md rather than a P-channel one, thesame transistor Md has its source connected to the organiclight-emitting diode OLED. As a result, the gate-to-source voltage Vgschanges with change of the same diode OLED over time.

This changes the drive current Id flowing through the organiclight-emitting diode OLED, thus changing the light emission brightnesseven if the data potential Vsig is at the predetermined level.

Further, a threshold voltage Vth and mobility μ of the drive transistorMd are different from one pixel circuit to another. This leads to avariation in the drain current Ids according to the equation shown inFIG. 4, thus changing the light emission brightness between differentpixels even if the supplied data potential Vsig is the same.

It should be noted that, in the equation shown in FIG. 4, referencenumeral Ids represents the current flowing between the drain and sourceof the drive transistor Md operating in the saturation region. Further,in the drive transistor Md, reference numeral Vth represents thethreshold voltage, μ the mobility, W the effective channel width(effective gate width), and L the effective channel length (effectivegate length). Still further, reference numeral Cox represents the unitgate capacitance of the drive transistor Md, namely, the sum of the gateoxide film capacitance per unit area and the fringing capacitancebetween the source (or drain) and gate.

The pixel circuit having the N-channel drive transistor Md isadvantageous in that it offers high driving capability and permitssimplification of the manufacturing process. To suppress the variationin the threshold voltage Vth and mobility p, however, the followingcorrection operations are demanded before the above-described lightemission control operation.

<Outline of the Corrections>

Although a detailed description will be given later, the gate-to-sourcevoltage Vgs of the drive transistor Md is maintained at the level of thethreshold voltage Vth by the holding capacitor Cs before the sampling.This preliminary operation is called the “threshold correction.”

A post-sampling data voltage Vin is added to the gate of the drivetransistor Md after the threshold correction. Therefore, thegate-to-source voltage Vgs changes to “Vth+Vin” and is maintained atthis level. The drive transistor Md turns on according to the magnitudeof the data voltage Vin. If the drive transistor Md does not easily turnon because of its large threshold voltage Vth, “Vth+Vin” is also large.In contrast, if the drive transistor Md easily turns on because of itssmall threshold voltage Vth, “Vth+Vin” is also small. This eliminatesthe impact of variation in the threshold voltage Vth from the drivecurrent, thus maintaining the drain current Ids (drive current Id)constant for the constant data voltage Vin.

Further, a “mobility correction” (driving capability correction to beprecise) is performed, for example, before the data sampling and afterthe threshold voltage correction.

The mobility correction changes the gate potential further according tothe current driving capability of the drive transistor Md with thevoltage “Vth+Vin” maintained constant. Although not illustrated in FIGS.2 and 3, the drive transistor Md has a path between its gate and sourceor drain. This path charges or discharges the holding capacitor with thecurrent supplied via the current channel of the same transistor Md. Themobility correction is performed by controlling whether or not to pass acurrent through this path.

Then, the organic light-emitting diode OLED emits light as it is drivenby this constant current.

<Pixel Circuit 3>

FIG. 5 illustrates a modification example of the pixel circuit 2 whichtakes into account the above charging-discharging path during themobility correction.

In the pixel circuit shown in FIG. 5, the holding capacitor Cs isconnected between the gate and source of the drive transistor Md ratherthan between the gate and drain thereof as illustrated in FIG. 3. Thepixel circuit in FIG. 5 is identical to that in FIG. 3 in otherconfiguration. It should be noted, however, that, here, power driving isachieved by driving the drain voltage of the drive transistor Md betweenhigh level (e.g., source voltage VDD) and low level (reference voltageVSS such as negative potential) using a power source drive pulse DS(i)(notation of a first scan signal VSCAN1(i) shown in FIG. 1 as a pulse).The same pulse DS(i) is supplied from the horizontal pixel line drivecircuit 41. Further, a video signal Ssig (data potential Vsig) issampled by the sampling transistor Ms using a write drive pulse WS(i)(notation of a second scan signal VSCAN2(i) shown in FIGS. 1 and 3). Thesame pulse WS(i) is supplied from the write signal scan circuit 42.

It should be noted that although the power driving of the pixel circuit3 is not limited to that shown in FIG. 5, we assume, for convenience ofconcrete description given later, that the power driving method shown inFIG. 5 is used.

Detailed Example of the Display Control

A description will be given of the operation of the circuit shown inFIG. 5 during the data write operation together with the thresholdvoltage and mobility correction operations. This series of operationsare referred to as the “display control.”

FIGS. 6A to 6I are timing diagrams illustrating the waveforms of varioussignals and voltages during the display control. Here, we assume thatdata is written on a row-by-row basis during the display control.Therefore, the first row having the pixel circuits 3(1,j) is the targetrow (display row). As a result, the second and third rows having thepixel circuits 3(2,j) and 3(3,j) are not the target rows (non-displayrows) at the point in time shown in FIGS. 6A to 6I. Data is written tothe display row through the display control which will be describedbelow. Then, the display row changes to the second row which undergoesthe same display control. The same display control is repeated on thethird, fourth and subsequent rows, thus allowing a screen to bedisplayed. After a screen is displayed, the display control is repeatedthe demanded number of times to display other screens in the samemanner.

FIG. 6A is a waveform diagram of the video signal Ssig.

FIGS. 6B and 6C are waveform diagrams of a write drive pulse WS(1) andpower drive pulse DS(1) supplied to the first row to which data is to bewritten. Similarly, FIGS. 6D and 6E are waveform diagrams of a writedrive pulse WS(2) and power drive pulse DS(2) supplied to the second rowto which data is not to be written. FIGS. 6F and 6G are waveformdiagrams of a write drive pulse WS(3) and power drive pulse DS(3)supplied to the third row to which data is not to be written.

FIG. 6H is a waveform diagram of the gate potential of the drivetransistor Md (potential of the control node NDc) in the pixel circuit3(1,j) in the first row to which data is to be written.

FIG. 6I is a waveform diagram of the source potential of the drivetransistor Md (anode potential of the organic light-emitting diode OLED)in the pixel circuit 3(1,j) in the first row to which data is to bewritten.

[Definitions of the Periods]

As illustrated at the bottom of FIG. 6I, FIGS. 6A to 6I show thewaveform diagrams over a time span about slightly more than four timesone horizontal period (1H) of the National Television system committee(NTSC) video signal standard. In the last horizontal period (1H), notonly the last or third threshold correction (VTC3) but also the mobilitycorrection and actual data write (W&μ) are performed continuously (mainoperation). The three horizontal periods ((1H)×3) preceding the lasthorizontal period (1H) are spent on performing the threshold correctiontwice in advance so that the correction progresses to a certain extent.This is done in consideration of the fact that the final thresholdcorrection may be too short to correct the threshold properly(preliminary operation).

With the display panel drive frequency already extremely high today, thedisplay control as shown in FIGS. 6A to 6I is unable to complete alloperations from the threshold voltage correction to data write within ashort one horizontal period (1H). Therefore, the threshold correction isperformed in several steps because of the lack of time available for thethreshold correction. It should be noted, however, that if onehorizontal period (1H) is enough for the main operation as in small- andmedium-sized display panels, one horizontal period (1H) may be enough toperform the initialization for the preliminary operation. Of course, thepreliminary operation may last for two horizontal periods (2H), or morethan four horizontal periods (4H).

When the main operation is conducted for a certain row, the preliminaryoperation can be performed for the next (and subsequent rows).Therefore, the length of the preliminary operation time hardly affectsthe display period as a whole. Rather, the preliminary operation shouldpreferably be conducted in a sufficient manner to ensure positivecompletion of the threshold voltage correction.

The above classification is based on a fixed measure, i.e., onehorizontal period (1H). However, it is also possible to functionallyunderstand the roughly four horizontal periods shown in FIG. 6I.

More specifically, as illustrated at the top of FIG. 6A, a lightemission period (LM0) for the screen preceding by one field (or frame)is followed by the “preliminary operation.” The preliminary operationincludes, in chronological order, a discharge period (D-CHG),initialization period (INT), first threshold correction period (VTC1),first standby period (WAT1), second threshold correction period (VTC2)and second standby period (WAT2). The preliminary operation is followedby the “main operation.” The main operation includes, in chronologicalorder, a third threshold correction period (VTC3), third standby period(WAT3) and writing and mobility correction period (W&μ). The mainoperation is followed by a light emission period (LM1) for the pixelcircuits 3(1,j) in the first row.

[Outline of the Drive Pulse]

Further, reference numerals T0 to T21 are shown as appropriate in thewaveform diagrams of FIGS. 6A to 6I to denote different times. Next, thevideo signal and drive pulse will be outlined with reference to thesetimes.

In the case of the write drive pulse WS(1) supplied to the first row,four sampling pulses (SP0 to SP3), which are inactive at low level andactive at high level, appear in a periodic manner. At this time, thesepulses (SP0 to SP3) occur at constant intervals throughout thepreliminary operation (time T0 to time T15) and main operation (aftertime T15). It should be noted, however, that the write drive pulse WS(1)in the main operation has a waveform in which a write pulse (WP) isadded after the fourth sampling pulse (SP3).

On the other hand, the video signal Ssig is supplied to the m (severalhundred to one thousand and several hundred) signal lines SIG(j) (referto FIGS. 1 and 5). The same signal Ssig is supplied simultaneously tothe m video signal lines Sig(j) in line sequential display. Asillustrated in FIG. 6A, the signal amplitude Vin reflecting the datavoltage obtained after the sampling of the video signal Ssig correspondsto the peak value of a video signal pulse (PP) which appears repeatedlyin the second half of one horizontal period (1H). This peak value isrelative to an offset potential (Vofs) which appears in the first halfof one horizontal period (1H). The signal amplitude Vin will behereinafter referred to as the data voltage Vin.

Of the several video signal pulses (PP) shown in FIG. 6A, the pulse(PPx) appearing during the main operation is essential for the firstrow. This pulse coincides in time with the write pulse (WP). The peakvalue of the video signal pulse (PPx) relative to the offset potential(Vofs) corresponds to the gray level to be displayed (written) shown inFIGS. 6A to 6I, i.e., the data voltage Vin. This gray level (=Vin) maybe the same for all pixels in the first row (for monochrome display).Normally, however, this level changes according to the gray level of thedisplay pixel row. FIGS. 6A to 6I are intended primarily to describe theoperation of one of the pixels in the first row. However, the driving ofother pixels in the same row is in itself controlled in parallel withthe driving of the single pixel illustrated in FIGS. 6A to 6I exceptthat the display gray level may be different between the pixels.

The power drive pulse DS(1) applied to the drain of the drive transistorMd (refer to FIG. 5) is maintained at inactive low level as illustratedin FIG. 6C from time T0 to immediately prior to the beginning (time T6)of the first threshold correction period (VTC1). The inactive low levelis, for example, the reference voltage VSS (e.g., negative voltage).Then, the power drive pulse DS(1) changes to active high level (e.g.,source voltage VDD) almost simultaneously with the beginning (time T6)of the first threshold correction period (VTC1). The same pulse DS(1) ismaintained at the source voltage VDD until the end of the light emissionperiod (LM1).

As illustrated in FIGS. 6D, 6E, 6F and 6G, the pulses WS and DSL areapplied to the pixel circuits 3(2,j) and 3(3,j) in the second and thirdrows respectively with a delay of one horizontal period.

More specifically, the first sampling pulse (SP0) for the initializationperiod (INT) is applied to the second row from time T5 to T7 duringwhich the second sampling pulse (SP1) for the first threshold correctionperiod (VTC1) is applied to the first row.

While this pulse is applied, that is, at time T6, the power drive pulseDS(1) for the first row changes to high level (source voltage VDD),which activates this pulse.

Then, the second sampling pulse (SP1) is applied to the second row witha delay of one horizontal period (1H) from that applied to the first rowfrom time T10 to T12 during which the third sampling pulse (SP2) for thesecond threshold correction period (VTC2) is applied to the first row.During the same time period, the first sampling pulse (SP0) is appliedto the third row with a delay of two horizontal periods ((1H)×2) fromthat applied to the first row.

While this pulse is applied, that is, at time T11, the power drive pulseDS(2) for the second row changes to high level (source voltage VDD),which activates this pulse.

Then, the third sampling pulse (SP2) is applied to the second row with adelay of one horizontal period (1H) from that applied to the first rowfrom time T15 to T17 during which the fourth sampling pulse (SP3) forthe third threshold correction period (VTC3) is applied to the firstrow. During the same time period, the second sampling pulse (SP1) isapplied to the third row with a delay of two horizontal periods ((1H)×2)from that applied to the first row.

While this pulse is applied, that is, at time T16, the power drive pulseDS(3) for the third row changes to high level (source voltage VDD),which activates this pulse.

Designing the pulse application timings as described above makes itpossible to perform, in parallel with the main operation of a given row,the preliminary operation of other rows whose main operation will beperformed one or more horizontal periods later. As far as the mainoperation is concerned, it is conducted on a row-by-row basis in aseamless manner. Therefore, there is no waste of time except for thefirst several horizontal periods.

Because the display screen normally includes several hundred to onethousand and several hundred rows, one to several horizontal periodsduring the display of one screen is negligibly small. Therefore, thereis substantially no time loss even when the threshold correction isperformed in several steps.

A description will be given next of the changes in potential of thesource and gate of the drive transistor Md shown in FIGS. 6H and 6I andthe operation associated with these changes when the pulses arecontrolled as described above. This description will be given for eachof the periods shown in FIG. 6A.

It should be noted that, here, reference will be made, as appropriate,to the explanatory diagram of the preliminary operation of the pixelcircuit 3(1,j) in the first row shown in FIGS. 7A to 9B, graph of thechange in the source potential Vs over time shown in FIG. 10,explanatory diagram of the main operation of the pixel circuit 3(1,j) inthe first row shown in FIGS. 11A to 11C and also to other drawings suchas FIG. 5.

[Light Emission Period for the Previous Screen (LM0)]

For the pixel circuit 3(1,j) in the first row, the write drive pulseWS(1) is at low level as illustrated in FIG. 6B during the lightemission period (LM0) for the screen preceding by one field or frameearlier than time T0 (hereinafter referred to as the previous screen).As a result, the sampling transistor Ms is off. At this time, on theother hand, the power drive pulse DS(1) is at the level of the sourcevoltage VDD as illustrated in FIG. 6C.

At this time, as illustrated in FIG. 7A, the organic light-emittingdiode OLED is emitting light according to a data voltage Vin0. The samevoltage Vin0 is held after being fed to the gate of the drive transistorMd by the data write operation for the previous screen. The drivetransistor Md is designed to operate in the saturation region.Therefore, the drive current Id (=Ids) flowing through the organiclight-emitting diode OLED takes on the value calculated by the equationshown in FIG. 4 according to the gate-to-source voltage Vgs of the drivetransistor Md held by the holding capacitor Cs.

[Discharge Period (D-CHG)]

The processes adapted to display a new screen by line sequential scanbegin from time T0 in FIGS. 6A to 6I.

At time T0, the horizontal pixel line drive circuit 41 (refer to FIG. 5)changes the power drive pulse DS(1) from the source voltage VDD to thereference voltage VSS as illustrated in FIG. 6C. In the drive transistorMd, the potential of the node which has been functioning as the drainuntil that time is sharply pulled down to the reference voltage VSS. Asa result, the relationship in potential between the source and drain isreversed. Therefore, the node which has been functioning as the drainserves as the source, and the node which has been functioning as thesource as the drain to discharge the charge from the drain (however,reference numeral Vs remains unchanged as the source potential).

Therefore, the drain current Ids flows in reverse direction through thedrive transistor Md as illustrated in FIG. 7B.

The period of time during which the current flows in reverse directionthrough the same transistor Md is written as the discharge period(D-CHG) in FIGS. 6A to 6I.

When the discharge period (D-CHG) begins, the source potential Vs (drainpotential in the practical operation) of the drive transistor Mddischarges sharply from time T0 as illustrated in FIG. 6I, causing thesame potential Vs to decline close to the reference voltage VSS.

At this time, if the reference voltage VSS is smaller than the sum of athreshold voltage Vth_oled. and a cathode potential Vcath of the organiclight-emitting diode OLED, i.e., VSS<Vth_oled.+Vcath, then the samediode OLED will stop emitting light.

It should be noted that the potential of the video signal Ssig is pulleddown from the data potential Vsig to the offset potential Vofs prior tothe end (time T1) of the discharge period (D-CHG), as illustrated inFIG. 6A.

At time T0, the sampling transistor Ms is off as illustrated in FIG. 7B,causing the control node NDc to float. As a result, the gate voltage Vgof the drive transistor Md declines from time T0 as illustrated in FIG.6H.

[Initialization Period (INT)]

Next, the write signal scan circuit 42 (refer to FIG. 5) changes thewrite drive pulse WS(1) from low to high level at time T1 as illustratedin FIG. 6B, thus supplying the first sampling pulse (SP0) to the gate ofthe sampling transistor Ms.

At time T1, the discharge period (D-CHG) ends, initiating theinitialization period.

In response to the application of the sampling pulse (SP0) at time T1,the sampling transistor Ms turns on as illustrated in FIG. 7C. Asdescribed earlier, the potential of the video signal Ssig is changed tothe offset potential Vofs by time T1. Therefore, the sampling transistorMs samples the offset potential Vofs of the video signal Ssig andtransfers the post-sampling offset potential Vofs to the gate of thedrive transistor Md.

This sampling operation causes the gate voltage Vg of the drivetransistor Md, which declined from time T0, to converge to the offsetpotential Vofs, as illustrated in FIG. 6H.

The sampling pulse (SP0) shown in FIG. 6B begins from time T1 and endsat time T2 when a sufficient time has elapsed for the convergence of thepotential, thus turning off the sampling transistor Ms. As a result, thegate of the drive transistor Md floats until time T5 when the samplingtransistor Ms turns on the next time.

The sampling transistor Ms is controlled to turn on again at time T5almost at the same time as the end of the first horizontal period (1H).Further, the same transistor Ms turns on again so that the video signalpulse (PP) fits into the first horizontal period (1H) (refer to FIGS. 6Aand 6B).

If this is viewed from the standpoint of the sampling pulse (SP0), theduration of the same pulse (SP0) (time T1 to T2) adapted to raise thewrite drive pulse WS(1) to high level is in the first half of thehorizontal period (1H) and falls within the period of time (time T0 toT3) during which the video signal Ssig is at the offset potential Vofs.

Then, at time T2, the sampling transistor Ms is turned off. With thesampling transistor Ms turned off, there is a wait until time T4 whenthe variation in potential of the signal line SIG(j) by the video signalpulse (PP) ends. Then, at time T5, the second sampling pulse (SP1) isactivated to sample the offset potential Vofs again.

This control prevents the data potential Vsig of the video signal Ssigto be erroneously sampled at time T5 when the second sampling pulse(SP1) is activated.

It should be noted that, as illustrated in FIG. 6H, the gate voltage Vgis already at the offset potential Vofs when the second sampling beginsat time T5. In general, therefore, the gate voltage Vg barely changesalthough the second sampling may make up for an extremely small losscaused, for example, by leak current.

Going slightly back to the description made on the time axis, thesampling transistor Ms turns on at time T1 as a result of theapplication of the first sampling pulse (SP0). When the gate voltage Vgof the drive transistor Md converges to the offset potential Vofs asillustrated in FIG. 6H, the voltage held by the holding capacitor Csdeclines to “Vofs−VSS” (FIG. 6I). This decline is caused by the factthat the discharge shown in FIG. 7B has pushed the source potential Vsdown to the reference voltage VSS and that the voltage held by theholding capacitor Cs is restricted by the gate voltage Vg relative tothe reference voltage VSS. That is, as the gate voltage Vg drops to theoffset potential Vofs, the voltage held by the holding capacitor Cs alsodrops and converges to “Vofs−VSS.” It should be noted that this heldvoltage “Vofs−VSS” is none other than the gate-to-source voltage Vgs.Unless the same voltage Vgs is greater than the threshold voltage Vth ofthe drive transistor Md, the threshold voltage correction operationcannot be performed later. As a result, the potential relationship isestablished so that “Vofs−VSS>Vth.”

As described above, the preparations for the threshold correctionoperation are completed by initializing the gate voltage Vg and sourcepotential Vs of the drive transistor Md.

[First Threshold Correction Period (VTC1)]

The sampling transistor Ms begins to sample the offset potential Vofsthe second time at time T5. Then, when the power drive pulse DS(1) risesfrom the VSS level to the VDD level at time T6, the initializationperiod (INT) ends, initiating the first threshold correction period(VTC1).

The sampling transistor Ms which is on is sampling the offset potentialVofs immediately prior to time T6, the beginning of the first thresholdcorrection period (VTC1). Therefore, the gate voltage Vg of the drivetransistor Md is electrically fixed at the constant offset potentialVofs.

In this condition, the horizontal pixel line drive circuit 41 (refer toFIG. 5) raises the power drive pulse DS(1) from low level (=VSS) to highlevel (=VDD) at time T6 as illustrated in FIG. 6C. From time T6 onward,the same circuit 41 maintains the potential of the power supply line tothe drive transistor Md at the source voltage VDD until the beginning ofthe processes for the next frame (or field).

As the power drive pulse DS(1) rises to high level, the “VDD−VSS”voltage is applied between the source and drain of the drive transistorMd. This causes the drain current Ids to flow through the drivetransistor Md.

The drain current Ids charges the source of the drive transistor Md,thus pushing up the source potential Vs as illustrated in FIG. 6I. As aresult, the gate-to-source voltage Vgs (voltage held by the holdingcapacitor Cs) of the drive transistor Md which has taken on the value“Vofs−VSS” up until that time declines gradually (FIGS. 6H and 6I).

At this time, the source of the drive transistor Md is not charged veryquickly by the drain current Ids. The reasons for this will be givenbelow with reference to FIG. 8A.

As illustrated in FIG. 8A, a gate bias voltage applied to the gate ofthe drive transistor Md is not very large because this voltage isrestricted by the offset voltage Vofs. Therefore, the drive transistorMd is only slightly on, that is, is on with only limited drivingcapability (first reason).

Further, although the drain current Ids flows into the holding capacitorCs, the same current Ids is also consumed to charge a capacitance Coled.of the organic light-emitting diode OLED. As a result, the sourcepotential Vs does not readily increase (second reason).

Still further, the sampling pulse (SP1) must be terminated at time T7which is before time T8 when the video signal Ssig changes to the datapotential Vsig the next time (refer to FIG. 6B). Therefore, the chargingtime of the source potential Vs is not sufficient (third reason).

Assuming that the second sampling pulse (SP1) shown in FIG. 6B can lastfor a sufficiently long time beyond time T7, the source potential Vs ofthe drive transistor Md (anode potential of the organic light-emittingdiode OLED) begins to increase from time T6 and continues to do so overtime and eventually converges to “Vofs−Vth” (curve CV shown by a dashedline in FIG. 10). That is, the source potential Vs should stopincreasing when the gate-to-source voltage Vgs (voltage held by theholding capacitor Cs) exactly matches the threshold voltage Vth of thedrive transistor Md.

[First Standby Period (WAT1)]

Practically, however, time T7 comes before the convergence point isreached. This terminates the duration of the sampling pulse (SP1), thusterminating the first threshold correction period (VTC1) and initiatingthe first standby period (WAT1).

More specifically, when the gate-to-source voltage Vgs of the drivetransistor Md becomes equal to Vx1 (>Vth), that is, when the sourcepotential Vs of the same transistor Md rises to “Vofs−Vx1” from thereference voltage VSS (at time T7), the first threshold correctionperiod (VTC1) ends. At this time (time T7), the voltage Vx1 is held bythe holding capacitor Cs.

When the first threshold correction period (VTC1) ends, the samplingtransistor Ms turns off. This places the gate of the drive transistorMd, which has been electrically fixed at the offset potential Vofs, in afloating state.

From time T7 onward, therefore, as the source potential Vs rises, thepotential of the gate in a floating state capacitively coupled to thesource will also rise (FIGS. 6H and 6I). As a result, in the presentexample, the source potential Vs increases toward the target convergencepoint “Vofs−Vth” (refer to FIG. 10) at the end (time T10) of the firststandby period (WAT1). On the other hand, the gate-to-source voltage Vgsremains unchanged as illustrated in FIGS. 6H and 6I.

As in the initialization period (INT) described earlier, it is necessaryto wait for the video signal pulse (PP) to elapse in the first standbyperiod (WAT1). Therefore, this period is called the “standby period” inthis respect. However, a relatively long standby period as that lastingfrom time T7 to T10 permits the gate voltage Vg to increase. Further,the gate-to-source voltage Vgs does not converge to the thresholdvoltage Vth as described above.

In FIG. 6H, the increment of the gate voltage Vg during the firststandby period (WAT1) is denoted by reference numeral Va1. Letting theincrement of the source potential Vs contributing to the increase in thegate voltage Vg through bootstrapping action via the couplingcapacitance (holding capacitor Cs) be also denoted by reference numeralVa1, the source potential Vs becomes equal to “Vofs−Vx1+Va1” at the end(time T10) of the first standby period (WAT1) (refer to FIG. 8B).

Therefore, it is necessary to bring the gate potential back to theoffset potential Vofs, i.e., the initialization level, and perform thethreshold voltage correction again.

[Second Threshold Correction Period (VTC2)]

In the operation example of the present embodiment, therefore, the sameprocesses as those performed during the first threshold correctionperiod (VTC1) and first standby period (WAT1) of the previous horizontalperiod (1H) (time T5 to T10) are performed during the next horizontalperiod (1H) (time T10 to T15). That is, the second threshold correctionperiod (VTC2) and second standby period (WAT2) are performed in the nexthorizontal period (1H).

However, the gate-to-source voltage Vgs (voltage held by the holdingcapacitor Cs) reduces to “Vx1” at time T10 when the second thresholdcorrection period (VTC2) begins. This “Vx1” is smaller than “Vofs-VSS”which is a relatively large value assumed by the gate-to-source voltageVgs (voltage held by the holding capacitor Cs) at time T5 when the firstthreshold correction period (VTC1) begins.

When the sampling transistor turns on at time T10 as the sampling pulse(SP2) rises as illustrated in FIG. 6B, the gate voltage Vg (=Vofs+Va1)of the drive transistor Md is connected to the signal line SIG(j) at alower potential (Vofs). This causes a current corresponding to thisdifference (Va1) to flow from the gate of the drive transistor Md to thesignal line SIG(j), forcing the gate voltage Vg down to the offsetpotential Vofs as illustrated in FIG. 8C.

The variation in potential (Va1) of the gate of the drive transistor Mdis fed to the source of the same transistor Md via the holding capacitorCs and a gate-to-source parasitic capacitance Cgs of the same transistorMd, thus pulling down the source potential Vs.

The decrement of the source potential Vs at this time is denoted byreference numeral g*Va1. Here, the capacitive coupling ratio g isexpressed as g=(Cgs+Cs)/(Cgs+Cs+Coled.) where Cgs represents thegate-to-source parasitic capacitance, (Cs) the same reference numeral asthe holding capacitor Cs and represents the capacitance thereof, andColed. represents the capacitance of the organic light-emitting diodeOLED. Therefore, the source potential Vs drops by “g*Va1” to“Vofs−Vx1+(1−g)Va1” from its immediately previous level or“Vofs−Vx1+Va1.”

The capacitive coupling ratio g takes on a value smaller than 1, as isclear from the definition equation. Therefore, the change “g*Va1” of thesource potential Vs is smaller than the change (Va1) of the gate voltageVg.

Here, if the gate-to-source voltage Vgs (=“Vx1−(1−g)Va1”) of the drivetransistor Md is greater than the threshold voltage Vth of the sametransistor Md, the drain current Ids flows as illustrated in FIG. 8C.The drain current Ids attempts to flow until the drive transistor Mdgoes into cutoff as a result of the source potential Vs of the drivetransistor Md becoming equal to “Vofs−Vth.” In the operation example ofthe present embodiment, however, the sampling pulse (SP2) ends at timeT12 when the gate-to-source voltage Vgs becomes equal to “Vx2” (whereVx2 is large enough to satisfy the condition Vx1>Vx2>Vth), asillustrated in FIGS. 6H and 6I. As a result, the sampling transistor Msturns off. The voltage held by the holding capacitor Cs at time T12 is“Vx2.”

[Second Standby Period (WAT2)]

The second standby period (WAT2) begins from time T12.

During the second standby period (WAT2), the sampling transistor Ms isoff, causing the gate voltage Vg to electrically float, as during theprevious first standby period (WAT1). As a result, as the sourcepotential Vs rises, the gate voltage Vg will also rise (refer to FIG.9A).

However, the effect of the increase in potential of the gate voltage Vg(bootstrapping effect) is not very large because the gate-to-sourcevoltage Vgs at the beginning of the standby period is close to thecontrol target “Vth.” As can be seen from time T12 to T15 in FIGS. 6Hand 6I, both the source potential Vs and gate voltage Vg increase onlyslightly.

More specifically, letting the increment of the source potential Vsresulting from the flow of the drain current during the second standbyperiod (WAT2) in FIG. 9A be denoted by reference numeral Va2, the sourcepotential Vs becomes equal to “Vofs−Vx2+Va2” at the end of the standbyperiod (time T15 in FIGS. 6A to 6I). This increase in the sourcepotential by “Va2” is transferred to the gate in a floating state viathe gate-to-source parasitic capacitance Cgs and holding capacitor Cs.As a result, the gate voltage Vg will also increase by the sameincrement or the potential Va2. It should be noted, however, that theincrement “Va2” of the potential of the gate voltage Vg is far smallerthan the increment “Va1” of the potential thereof during the firststandby period (WAT1) as illustrated in FIG. 6H.

[Third Threshold Correction Period (VTC3)]

The “main operation” begins from time T15, initiating the thirdthreshold correction period (VTC3).

The same processes as those performed during the second thresholdcorrection period (VTC2) are performed during time T15 to T17 the thirdthreshold correction period (VTC3).

However, the gate-to-source voltage Vgs (voltage held by the holdingcapacitor Cs) reduces to “Vx2” at time T15 when the third thresholdcorrection period (VTC3) begins. This “Vx2” is even smaller than “Vx1”which is a relatively large value assumed by the gate-to-source voltageVgs (voltage held by the holding capacitor Cs) at time T10 when thesecond threshold correction period (VTC2) begins.

The description of the basic operation will be omitted to avoidredundancy. The description of the second threshold correction period(VTC2) is applicable to the third threshold correction period (VTC3) byreplacing “Va1” with “Va2” and “Vx1” with Vx2.” This is also obviousfrom the comparison between FIG. 8C and FIG. 9B.

It should be noted that the third threshold correction period (VTC3)differs from the second one (VTC2) in that the gate-to-source voltageVgs (voltage held by the holding capacitor Cs) becomes equal to thethreshold voltage Vth by time T17 when the third threshold correctionperiod (VTC3) ends, as illustrated in FIGS. 6H and 6I. Therefore, thedrive transistor Md goes into cutoff when the gate-to-source voltage Vgsbecomes equal to the threshold voltage Vth. From this moment onward, thedrain current Ids will not flow. At this time, the source potential Vsof the drive transistor Md is “Vofs−Vth.”

As described above, the threshold correction performed a plurality oftimes (three times in the present example) with a standby periodprovided therebetween permits the voltage held by the holding capacitorCs to converge in a stepped manner. In the course of the convergence,the held voltage remains constant during the standby periods andeventually converges to the threshold voltage Vth.

Here, assuming that the gate-to-source voltage of the drive transistorincreases by “Vin,” then the same voltage is equal to “Vin+Vth.” Weconsider two drive transistors, one having the large threshold voltageVth and another having the small threshold voltage Vth.

The former with the large threshold voltage Vth has the largegate-to-source voltage commensurate with the large threshold voltageVth. In contrast, the latter with the small threshold voltage Vth hasthe small gate-to-source voltage commensurate with the small thresholdvoltage Vth. Therefore, as far as the threshold voltage Vth isconcerned, it is possible to pass the same amount of the drain currentIds through the drive transistor for the same data potential Vin bycanceling the variation in the threshold voltage Vth using the thresholdvoltage correction operation.

During the three threshold correction periods, namely, the first, secondand third threshold correction periods (VTC1), (VTC2) and (VTC3), it isnecessary to ensure that the drain current Ids is wholly consumed for itto flow into one of the electrodes of the holding capacitor Cs, i.e.,one of the electrodes of the capacitance Coled. of the organiclight-emitting diode OLED, so that the same diode OLED does not turn on.If the anode voltage of the same diode OLED is denoted by referencenumeral Voled., the threshold voltage thereof by reference numeralVth_oled., and the cathode voltage thereof by reference numeral Vcath,the equation “Voled.≦Vcath+Vth_oled.” must always hold in order for thesame diode OLED to remain off.

Assuming here that the cathode potential Vcath of the organiclight-emitting diode OLED is constant at the reference voltage VSS(e.g., ground voltage GND), the above equation can hold at all times ifthe threshold voltage Vth_oled. is extremely large. However, thethreshold voltage Vth_oled. is determined by the manufacturingconditions of the organic light-emitting diode OLED. Further, the samevoltage Vth_oled. cannot be increased excessively to achieve efficientlight emission at low voltage. Therefore, the organic light-emittingdiode OLED should preferably be reverse-biased by setting the cathodepotential Vcath larger than the reference voltage VSS until the threethreshold correction periods and the mobility correction period, whichwill be described below, ends.

[Third Standby Period (WAT3)]

A description has been given above of the threshold voltage correction.In the present operation example, the threshold voltage correction isfollowed by a standby period for the writing and mobility correction(third standby period (WAT3)). Unlike the first and second standbyperiods (WAT1) and (WAT2), the third standby period (WAT3) is a shortperiod of time designed simply to prevent the erroneous sampling of thevideo signal Ssig at an unstable potential during the writing andmobility correction performed thereafter.

As illustrated in FIG. 6B, the third standby period (WAT3) begins attime T17 when the sampling pulse (SP3) changes from high to low level.

In the third standby period (WAT3), the video signal pulse (PPx) havingthe data potential Vsig to be displayed by the pixel circuit 3(1,j) asillustrated in FIG. 6A is supplied to the signal line SIG(j) as thevideo signal Ssig at time T18 during this period (refer to FIG. 11A). Inthe video signal Ssig, the difference between the data potential Vsigand offset potential Vofs corresponds to the data voltage Vin for thegray level to be displayed by the pixel circuit. That is, the datapotential Vsig is equal to “Vofs+Vin.”

The third standby period (WAT3) ends at time T19 when the video signalSsig is constant at the data potential Vsig following the change inpotential at time 18.

[Writing and Mobility Correction Period (W&μ)]

The writing and mobility correction period (W&μ) begins from time T19.

As illustrated in FIG. 6B, the write pulse (WP) is supplied to the gateof the sampling transistor Ms at time T19 during the application of thevideo signal pulse (PPx) for the main operation. This turns on thesampling transistor Ms, causing the difference between the datapotential Vsig (=Vin+Vofs) of the signal line SIG(j) and the gatevoltage Vg (=Vofs), i.e., the data voltage Vin, to be fed to the gate ofthe drive transistor Md, as illustrated in FIG. 11B. As a result, thegate voltage Vg is equal to “Vofs+Vin.”

As the gate voltage Vg rises by the data voltage Vin, the source voltageVs will also rise. At this time, the data voltage Vin is not transferred“as-is” to the source potential Vs. Instead, the source potential Vswill rise only by the change of the data voltage Vin proportional to thecapacitive coupling ratio g, i.e., “g*Vin.” Therefore, the sourcepotential Vs after the change is equal to “Vofs−Vth+g*Vin.” As a result,the gate-to-source voltage Vgs of the drive transistor Md is equal to“(1−g)Vin+Vth.”

Here, a description will be given of the variation due to the mobilityμ.

In the three threshold voltage corrections performed up to this point,the drain current Ids contains, in fact, an error resulting from themobility μ each time this current flows. However, the error caused bythe mobility μ was not treated as problematic because the variation inthe threshold voltage Vth was large. At this time, the description wasgiven by writing the voltages simply as “Va1” and “Va2” to represent theresults rather than using the capacitive coupling ratio g. This was doneto avoid complication associated with describing the variation in themobility.

As already explained, on the other hand, the threshold voltage Vth isheld by the holding capacitor Cs after a threshold voltage correctionperformed in a strict manner. If the drive transistor Md is turned onthereafter, the drain current Ids will remain unchanged irrespective ofthe magnitude of the threshold voltage Vth. Therefore, assuming that thevoltage held by the holding capacitor Cs (gate-to-source voltage Vgs)changes due to the drive current Id at the time of the conduction of thedrive transistor Md after the threshold voltage correction, this changeΔV (positive or negative) reflects not only the variation in themobility μ of the drive transistor Md, and more precisely, the mobilitywhich, in a strict sense, is a physical parameter of the semiconductormaterial, but also the comprehensive variation in those factorsaffecting the current driving capability in terms of transistorstructure or manufacturing process.

Going back to the description of the operation in consideration of theabove, when the data voltage Vin is added to the gate potential Vg afterthe sampling transistor Ms has turned on in FIG. 11B, the drivetransistor Md attempts to pass the drain current Ids, commensurate inmagnitude with the data voltage Vin (gray level), between the drain andsource. At this time, the drain current Ids varies according to themobility μ. As a result, the source potential Vs is given by“Vofs−Vth+g*Vin+ΔV,” which is the sum of “Vofs−Vth+g*Vin” and the changeΔV resulting from the mobility μ.

At this time, in order for the organic light-emitting diode OLED not toemit light, it is only necessary to set the cathode potential Vcath inadvance according, for example, to the data voltage Vin and capacitivecoupling ratio g so that the equationVs(=Vofs−Vth+g*Vin+ΔV)<Vth_oled.+Vcath is satisfied.

Setting the cathode potential Vcath in advance as described abovereverse-biases the organic light-emitting diode OLED, bringing the samediode OLED into a high impedance state. As a result, the organiclight-emitting diode OLED does not emit light and exhibits a simplecapacitive characteristic rather than a diode characteristic.

At this time, so long as the above equation is satisfied, the sourcepotential Vs will not exceed the sum of the threshold voltage Vth_oled.and cathode potential Vcath of the organic light-emitting diode OLED.Therefore, the drain current Ids (drive current Id) is used to charge acombined capacitance C=Cs+Coled.+Cgs which is the sum of threecapacitance values. These are the capacitance of the holding capacitorCs (denoted by the same reference numeral Cs), that of the equivalentcapacitance of the organic light-emitting diode OLED (denoted by thesame reference numeral Coled. as the parasitic capacitance) when thesame diode OLED is reverse-biased and that of a parasitic capacitance(denoted by Cgs) existing between the gate and source of the drivetransistor Md. This causes the source potential Vs of the drivetransistor Md to rise. At this time, the threshold voltage correctionoperation of the drive transistor Md is already complete. Therefore, thedrain current Ids flowing through the same transistor Md reflects themobility μ.

As shown in the equation (1−g)Vin+Vth−ΔV in FIGS. 6H and 6I, as far asthe gate-to-source voltage Vgs held by the holding capacitor Cs isconcerned, the change ΔV added to the source potential Vs is subtractedfrom the gate-to-source voltage Vgs (=(1−g)Vin+Vth) after the thresholdvoltage correction. Therefore, the change ΔV is held by the holdingcapacitor Cs so that a negative feedback is applied. As a result, thechange ΔV will also be hereinafter referred to as a “negative feedbackamount.”

The negative feedback amount ΔV can be expressed by the approximationequation ΔV=t*Ids/Coled. because the equation Coled. >>Cs+Cgs holds whenthe organic light-emitting diode OLED is reverse-biased. It is clearfrom this approximation equation that the change ΔV is a parameter whichchanges in proportion to the change of the drain current Ids.

From the approximation equation of the feedback amount ΔV, the sameamount ΔV added to the source potential Vs is dependent upon themagnitude of the drain current Ids (this magnitude has a positivecorrelation with the magnitude of the data voltage Vin, i.e., the graylevel) and the period of time during which the drain current Ids flows,i.e., time (t) from time T19 to time T20 demanded for the mobilitycorrection shown in FIG. 6B. That is, the larger the gray level and thelonger the time (t), the larger the negative feedback amount ΔV.

Therefore, the mobility correction time (t) need not always be constant.Rather, it may be more appropriate to adjust the mobility correctiontime (t) according to the drain current Ids (gray level). For example,when the gray level is almost white with the drain current Ids beinglarge, the mobility correction time (t) should be short. In contrast,when the gray level is almost black with the drain current Ids beingsmall, the mobility correction time (t) should be long. This automaticadjustment of the mobility correction time according to the gray levelcan be implemented by providing this functionality, for example, in thewrite signal scan circuit 42 in advance.

[Light Emission Period (LM1)]

The writing and mobility correction period (W&μ) ends at time T20,initiating the light emission period (LM1).

The write pulse (WP) ends at time T20, turning off the samplingtransistor Ms and causing the gate of the drive transistor Md toelectrically float.

Incidentally, in the writing and mobility correction period (W&μ) priorto the light emission period (LM1), the drive transistor Md may notalways be able to pass the drain current Ids commensurate with the datavoltage Vin despite its attempt to do so. The reason for this is asfollows. That is, the gate voltage Vg of the drive transistor Md isfixed at Vofs+Vin if the current level (Id) flowing through the organiclight-emitting diode OLED is considerably smaller than that (Ids)flowing through the drive transistor Md because the sampling transistorMs is on. The source potential Vs attempts to converge to the potential(Vofs+Vin−Vth) which is lower by the threshold voltage Vth fromVofs+Vin. Therefore, no matter how long the mobility correction time (t)is extended, the source potential Vs will not exceed the aboveconvergence point. The mobility should be corrected by monitoring thedifference in the mobility μ based on the difference in time demandedfor the convergence. Therefore, even if the data voltage Vin close towhite that has the maximum brightness is supplied, the end point of themobility correction time (t) is determined before the convergence isachieved.

When the gate of the drive transistor Md floats after the light emissionperiod (LM1) has begun, the source potential Vs of the same transistorMd is allowed to rise further. Therefore, the drive transistor Md actsto pass the drive current Id commensurate with the supplied data voltageVin.

This causes the source potential Vs (anode potential of the organiclight-emitting diode OLED) to rise. After a while, the organiclight-emitting diode OLED is no longer reverse-biased. As a result, thedrain current Ids begins to flow through the same diode OLED as thedrive current Id as illustrated in FIG. 11C, causing the same diode OLEDto emit light. Shortly after the light emission begins, the drivetransistor Md is saturated with the drain current Ids commensurate withthe supplied data voltage Vin. When the same current Ids (=Id) isbrought to a constant level, the organic light-emitting diode OLED willemit light at the brightness commensurate with the data voltage Vin.

The increase in the anode potential of the organic light-emitting diodeOLED taking place from the beginning of the light emission period (LM1)to when the brightness is brought to a constant level is none other thanthe increase in the source potential Vs of the drive transistor Md. Thisincrease in the source potential Vs will be denoted by reference numeralΔVoled. to represent the increment of the anode voltage Voled. of theorganic light-emitting diode OLED. The source potential Vs of the drivetransistor Md becomes equal to “Vofs-Vth+g*Vin+ΔV+ΔVoled (refer to FIG.6I).

On the other hand, the gate potential Vg increases by the incrementΔVoled. as does the source potential Vs as illustrated in FIG. 6Hbecause the gate is floating. As the drain current Ids saturates, thesource potential Vs will also saturate, causing the gate potential Vg tosaturate.

As a result, the gate-to-source voltage Vgs (voltage held by the holdingcapacitor Cs) is maintained at the level during the mobility correction(“(1−g) Vin+Vth−ΔV”) throughout the light emission period (LM1).

During the light emission period (LM1), the drive transistor Mdfunctions as a constant current source. As a result, the I-Vcharacteristic of the organic light-emitting diode OLED may change overtime, changing the source potential Vs of the drive transistor Md.

However, the voltage held by the holding capacitor Cs is maintained at(1−g)Vin+Vth−ΔV, irrespective of whether the I-V characteristic of theorganic light-emitting diode OLED changes over time. The voltage held bythe holding capacitor Cs contains two components, (+Vth) adapted tocorrect the threshold voltage Vth of the drive transistor Md and (−ΔV)adapted to correct the variation due to the mobility μ. Therefore, evenif there is a variation in the threshold voltage Vth or mobility μbetween different pixels, the drain current Ids of the drive transistorMd, i.e., the drive current Id of the organic light-emitting diode OLED,will remain constant.

More specifically, the larger the threshold voltage Vth, the more thedrive transistor Md reduces the source potential Vs using the thresholdvoltage correction component (+Vth) contained in the held voltage, thusincreasing the source-to-drain voltage so that the drain current Ids(drive current Id) flows in a larger amount. Therefore, the draincurrent Ids remains constant even in the event of a change in thethreshold voltage Vth.

On the other hand, if the change ΔV is small because of the smallmobility μ, the voltage held by the holding capacitor Cs will declineonly to a small extent thanks to the mobility correction component (−ΔV)contained therein. This provides a relatively large source-to-drainvoltage. As a result, the drive transistor Md operates in such a manneras to pass the drain current Ids (drive current Id) in a larger amount.Therefore, the drain current Ids remains constant even in the event of achange in the mobility μ.

As described above, the light emission brightness of the organiclight-emitting diode OLED is maintained constant so long as the datavoltage Vin is the same even in the event of a variation in thethreshold voltage Vth or mobility μ between different pixels, andfurther irrespective of the change in I-V characteristic of the organiclight-emitting diode OLED over time.

Examples of Planar and Sectional Structure

Based on the above block and circuit configurations, a description willbe given next of the planar pattern and sectional structure of the pixelcircuit with reference to the accompanying drawings.

FIGS. 12A and 12B illustrate a planar pattern of the pixel circuit3(i,j) in the ith row and jth column. FIG. 12B is a plan view with thecathode electrode on the uppermost layer (formed over the entiresurface) omitted. FIG. 12A is a plan view midway during the manufacturewith the electrodes and organic multi-layered film of the organiclight-emitting diode OLED including the cathode electrode on theuppermost layer (formed over the entire surface) omitted.

FIG. 13A is a schematic sectional view taken on line A-A in FIG. 12A.FIG. 13B is a schematic sectional view taken on line B-B in FIGS. 12Aand 12B.

In FIGS. 13A and 13B, an underlying layer 10 (type of insulating layer)is formed directly on an unshown substrate made, for example, of glassor indirectly via other film.

In the sectional view shown in FIG. 13B, a gate electrode 11A is formedon the underlying layer 10. The gate electrode 11A includes a given gatemetal layer (GM) and contains a high-fusion-point metal such asmolybdenum (Mo). The sectional view in FIG. 13B shows where the drivetransistor Md illustrated, for example, in FIG. 5 is formed. Asillustrated in FIG. 12A, a gate electrode 11D, slightly different insize from the gate electrode 11A, is similarly formed where the samplingtransistor Ms is formed.

In the sectional view shown in FIG. 13A, on the other hand, two layersare formed on the underlying layer 10, namely, first and second layers(patterns) which include the gate metal layer (GM) provided at the samehierarchical level and made of the same material as the gate electrode11A and which contain a high-fusion-point metal wiring (hereinafterreferred simply to as the first and second high-fusion-point metalwiring layers). The first high-fusion-point metal wiring layer (pattern)11B and second high-fusion-point metal wiring layer (pattern) 11C arespaced apart within the pixel but continuous between the adjacentpixels, as illustrated in FIG. 12A. That is, the first high-fusion-pointmetal wiring layer 11B shown in FIG. 12A is connected to the secondhigh-fusion-point metal wiring layer 11C (not shown) in the otherunshown pixel which is continuous as a pattern on one side in the columndirection (downward from FIG. 12A). Similarly, the secondhigh-fusion-point metal wiring layer 11C shown in FIG. 12A is connectedto the first high-fusion-point metal wiring layer 11B (not shown) in theother unshown pixel which is continuous as a pattern on the other sidein the column direction (upward from FIG. 12A).

A gate insulating film 12 is formed over the entire surface of theunderlying layer 10 to cover the surfaces of the gate electrode 11A(FIG. 13B) and the first and second high-fusion-point metal wiringlayers 11B and 11C (FIG. 13A).

In the sectional view shown in FIG. 13B, a TFT layer 13A of the drivetransistor Md is formed on the gate insulating film 12. The TFT layer13A is made, for example, of amorphous silicon (or polysilicon forP-channel TFT). As illustrated in FIG. 12A, a TFT layer 13B of thesampling transistor Ms is formed similarly although different in sizefrom the TFT layer 13A. The TFT layer 13A shown in FIG. 13B is dopedwith an impurity of opposite conductivity type, thus forming source (S)and drain (D) regions which are separated from each other. The sameholds true for the TFT layer 13B.

In the sectional view shown in FIG. 13A, a plurality of contact holes orcontact holes 12A and 12B are formed on the end portion of the firsthigh-fusion-point metal wiring layer 11B in the gate insulating film 12.Similarly, a plurality of contact holes or contact holes 12C and 12D areformed on the end portion of the second high-fusion-point metal wiringlayer 11C in the gate insulating film 12.

A total of the four contact holes 12A to 12D or two holes for each ofthe wiring connection sections serve as first contact holes (1CH)adapted to connect together the high-fusion-point metal wiring layer andits overlying layer.

More specifically, the first high-fusion-point metal wiring layer 11Bhas its end portion connected to one of the end portions of an overlyingwiring layer 14B via the contact holes 12A and 12B. The overlying wiringlayer (pattern) 14B is provided on the gate insulating film 12 and made,for example, of aluminum (AL). Further, the second high-fusion-pointmetal wiring layer 11C has its end portion connected to the other endportion of the overlying wiring layer 14B provided on the gateinsulating film 12 via the contact holes 12C and 12D.

A supply line of the source voltage VDD (hereinafter referred to as thesource voltage supply line VDDL) is provided above the firsthigh-fusion-point metal wiring layer 11B. The source voltage supply lineVDDL is insulated from the same layer 11B via the gate insulating film12 and separated from the overlying wiring layer 14B by a pattern. Thesame line VDDL is connected to the horizontal pixel line drive circuit41 shown in FIG. 5 and is designed to alternately apply the sourcevoltage VDD and reference voltage VSS to the drain of the drivetransistor Md. Therefore, a branch (denoted by the same referencenumeral VDDL) of the source voltage supply line VDDL is inlow-resistance electrical contact with the region which will serve asthe drain (D) of the TFT layer 13A. On the other hand, an upperelectrode layer (pattern) 14D of the holding capacitor Cs is inelectrical contact with the region which will serve as the source (S) ofthe drive transistor Md. The upper electrode layer 14D is provided atthe same hierarchical level and made of the same material (aluminum AL)as the source voltage supply line VDDL. As illustrated in FIG. 12A, thesame layer 14D overlaps the lower electrode layer of the holdingcapacitor Cs which is continuous from the gate electrode 11A. Thisportion forms the holding capacitor Cs having a MIS(Metal-Insulator-Semiconductor) structure.

In FIG. 13B, a control line SAML of the sampling transistor Ms isprovided above the second high-fusion-point metal wiring layer 11C. Thecontrol line SAML is insulated from the same layer 11C via the gateinsulating film 12 and separated from the overlying wiring layer 14B bya pattern. The same line SAML is connected to the write signal scancircuit 42 shown in FIG. 5 and is designed to apply the write drivepulse WS(i) to the gate of the sampling transistor Ms. As illustrated inFIG. 12A, therefore, the control line SAML is connected to the gateelectrode 11D of the sampling transistor Ms in the underlying layer viaa contact hole 12E which is one of the first contact holes (1CH).

The control line SAML is disposed long in the row direction in parallelwith the source voltage supply line VDDL. The signal line SIG(j) has astructure in which the second high-fusion-point metal wiring layer 11Cserves as a lower bridge at the intersection with the control line SAML(referred to as the lower bridge structure in the presentspecification). Similarly, the signal line SIG(j) has a structure inwhich the first high-fusion-point metal wiring layer 11B serves as alower bridge at the intersection with the source voltage supply lineVDDL (lower bridge structure).

It should be noted that the overlying wiring layer 14B is connected, onthe pattern, to the drain side of the TFT layer 13B of the samplingtransistor Ms and that an in-cell wiring 14E, made of aluminum (AL) andmaking up part of the control node NDc of the drive transistor Md shownin FIG. 5, is connected to the source side thereof. The in-cell wiring14E is electrically connected to the lower electrode layer of theholding capacitor Cs in the underlying layer via a contact hole 12Fwhich is one of the first contact holes (1CH).

A planarizing film 15 is formed over the entire surface to bury thealuminum (AL) wirings formed as described above, i.e., the sourcevoltage supply line VDDL, control line SAML, overlying wiring layer 14Band upper electrode layer 14D, and planarize the surface by removing thelevel differences therebetween (refer to FIG. 13B).

As shown in the sectional view in FIG. 13B, an anode contact 15A isformed in a portion of the planarizing film 15 on the upper electrodelayer 14D by filling a second contact hole (2CH), formed in theplanarizing film 15, with a conductive material.

Then, an anode electrode (AE), a protective film 16, an organicmulti-layered film (OML) and a cathode electrode (CE) are deposited inthis order, thus forming the organic light-emitting diode OLED. Theanode electrode (AE) is formed on the planarizing film 15 and in contactwith the end surface of the anode contact 15A. The protective film 16 isformed on the anode electrode (AE) and has an opening portion 16A onesize smaller than the anode contact 15A. The organic multi-layered film(OML) covers the protective film 16. The cathode electrode (CE) isformed in the form of a blanket over the entire surface of the areaoccupied by the pixel. As illustrated in FIG. 12B, the protective film16 having the opening portion 16A one size smaller than the anodeelectrode (AE) is formed on the cathode electrode (CE).

In the signal line SIG(j) having the pattern and sectional structure asdescribed above, a plurality of contact holes such as the contact holes12A to 12D are formed for each wiring connection section. That is, thecontact holes 12A and 12B are formed for the wiring connection sectionbetween the first high-fusion-point metal wiring layer 11B and overlyingwiring layer 14B forming the lower bridge structure. The contact holes12C and 12D are formed for the wiring connection section between thesecond high-fusion-point metal wiring layer 11C and overlying wiringlayer 14B forming the lower bridge structure. This provides a smallerdelay of the video signal Ssig supplied from the H scanner 5 shown inFIG. 5 to the signal line SIG(j) than connection by a single contacthole.

Further, the control line SAML is the single layer wiring 14C made oflow-resistance aluminum (AL) and has no contacts midway along thewiring. This provides a smaller delay of the write drive pulse WS(i)supplied from the write signal scan circuit 42 shown in FIG. 5 to thecontrol line SAML than if the control line SAML were wholly or partlyformed with the gate metal (GM).

In designing the sampling timing of the data potential Vsig using thesampling transistor Ms, therefore, the delays of the two signals arereduced. This provides a wider sampling margin for the variation inlevel of the data potential Vsig, thus ensuring greater resistance todisplay quality degradation. In other words, the present embodiment isadvantageous in that the wider sampling margin provides greater leewayfor achieving larger screen size and higher definition (improved drivefrequency).

Any number of contact holes may be provided for each wiring connectionsection so long as there are at least two holes. It should be noted,however, that this number should preferably be greater than the maximumnumber of contact holes provided for each of the other connectionsections in the pixel including the connection section adapted toconnect the control line SAML to the control node (gate) of the samplingtransistor Ms. That is, in the case of the example shown in FIGS. 13Aand 13B, there is one contact provided for each “connection section,”namely, the contact hole 12E, contact hole 12F and anode contact 15A. Incontrast, there are two contacts provided for each wiring connectionsection, namely, a pair of the contact holes 12A and 12B and a pair ofthe contact holes 12C and 12D.

When the number of the contact holes is increased for each of the wiringconnection sections, only a given number of the contact holes areprovided if the waveform of the pulse of the video signal Ssig havingits active level at the data potential Vsig no longer shows any changein potential with the given number of the contact holes or moreprovided. More specifically, in the above example, there are two contactholes for each of the wiring connection sections, namely, a pair of thecontact holes 12A and 12B and a pair of the contact holes 12C and 12D.Now, we assume that this number is increased to 3, 4 and so on. When thenumber of contact holes is increased, the video signal pulse (PP) or(PPx) of the video signal Ssig may be regarded as showing substantiallyno change in waveform with a given number of the contact holes or moreprovided. The term “showing substantially no change in waveform” may bedefined arbitrarily. For example, however, assuming that the pulse ismonitored when it is 90% of its peak value, and if this monitored changepoint of the waveform falls within a given tolerance in all the pixelsin the intended row of the pixel array 2, the signal may be regarded as“showing substantially no change in waveform.” If the fact that there issubstantially no change in waveform is detected, it is only necessary toestimate the minimum number of contacts at the time of detection anddetermine this minimum number (2 or more) as the number of contacts perwiring connection section.

Still further, as illustrated in the sectional view of FIG. 14 takenalong line A-A in FIG. 12A as with that of FIG. 13A, the first andsecond high-fusion-point metal wiring layers 11B and 11C are formed tobe connected together on the pattern of the gate metal layer (GM). Then,the overlying wiring layer 14B is formed to the maximum length with anecessary spacing width ‘d’ provided on both ends. Prior to thisformation, as many first contacts (1CH) as can be disposed along themaximum length of the overlying wiring layer 14B or 11 contact holes 12Ato 12K in this case are formed in advance in the gate insulating film12.

In such a structure, even if a high-resistance state arises as a resultof the aluminum coming close to breaking for unknown reason, there is apath connection near the high resistance portion via the portion of thehigh-fusion-point metal wiring layer in the underlying layer. Therefore,even if the aluminum (AL) increases in resistance, a repair route forthe wiring path will be formed to ensure that the wiring resistance doesnot increase to the extent possible because the aluminum is shortalthough high in resistance. This provides a wiring structureunsusceptible to the increased resistance. It should be noted that thisstructure is more preferred for its smaller wiring resistance than thatshown in FIG. 13A even in the absence of any break.

The increased resistance of the signal line SIG(j) is likely to lead toserious display defects in the form of line defects rather than pointdefects. However, the above embodiment is significantly advantageous inthat it can prevent display quality degradation by preventing linedefects.

The present application contains subject matter related to thatdisclosed in Japanese priority Patent Application JP 2008-126118 filedin the Japan Patent Office on May 13, 2008, the entire content of whichis hereby incorporated by reference.

It should be understood by those skilled in the art that variousmodifications, combinations, sub-combinations and alterations may occurdepending on design requirements and other factors insofar as they arewithin the scope of the appended claims or the equivalents thereof.

What is claimed is:
 1. An apparatus comprising: a plurality of datalines, at least one of the plurality of data lines being configured tosupply a data signal to a pixel capacitor via a first transistor; and aplurality of power supply lines, at least one of the plurality of powersupply lines being configured to supply a driving current to a lightemitting element via a second transistor, wherein, the at least one ofthe plurality of data lines comprises a lower wiring portion includingMolybdenum (Mo) and an upper wiring portion including Aluminum (Al), theupper wiring portion being connected to the lower wiring portion via aplurality of contact holes, the at least one of the plurality of powersupply lines includes Aluminum (Al), the upper wiring portion is higherhierarchical level than the lower wiring portion, the at least one ofthe plurality of data lines extends in a column direction, the pluralityof contact holes are arranged along the column direction, and the lowerwiring portion crosses the at least one of the plurality of power supplylines.